In many applications such as motor control, solenoid valve control, communication infrastructure, and power management, current detection is a key function necessary for precision closed-loop control. It can be found in automotive and industrial applications where safety is paramount to handheld devices where power and efficiency are paramount. Using precision current monitoring, designers can obtain key instantaneous information, such as motor torque (based on motor current), DC/DC converter efficiency, base station LDMOS (laterally diffused MOS) power transistor bias current, or short-circuit to ground, etc. Diagnostic information.
In order to understand the important trade-offs, choices, and challenges that system designers face when choosing the most accurate and cost-effective current sensor for the circuit board, we will carefully discuss the LDMOS bias current monitoring of cellular base station power amplifiers and other related applications. Current detection in.
Current monitoring is indispensable in base station power amplifiers, especially in 3G and LTE with more complex modulation methods. The peak-to-average power ratio ranges from 3GW-CDMA’s 3.5dB (approximately 2.2 to 1) to LTEOFDM’s 8.5dB (Approximately 7.1 to 1) range, and the peak-to-average power ratio of most commonly used 2G single-carrier GSM is 3dB (approximately 2 to 1). One of the functions of the control loop is to monitor the LDMOS bias current so that the LDMOS bias can be correctly modulated for a given power output. Normally, this DC bias current has a wide dynamic range, depending on operating conditions, maximum value, or off-peak operation. For designers, this means that a precision current sensor is needed to monitor 50mA (or as low as 15mA)1To a current in the range of 20A, and the drain of the LDMOS is biased to a high voltage in the range of 28V to 60V. If a shunt resistor is used to monitor this current, the designer can only use a very small shunt resistor, otherwise when the LDMOS current is 20A, its power consumption will be very large. For example, at maximum current, even a 10mΩ shunt resistor will consume 4W of power.
Although there are shunt resistors that can withstand this power, the circuit board may require lower power consumption. However, if such a low resistance value is selected, at low currents (such as 50mA), the voltage on the 10mΩ shunt resistor will be extremely small (500μV), and it is difficult to use a circuit that must also withstand high common-mode voltages for precision monitoring.
This article will focus on current detection solutions that can accurately monitor a wide range of DC currents under high common-mode voltages. At the same time, special attention will be paid to the important parameter of temperature performance, which is often difficult to calibrate, but it must be treated with caution in outdoor applications of power amplifiers. This article will introduce three alternative solutions in descending order of design complexity, which can provide feasible high-precision, high-resolution current detection for a variety of different applications.
- Use discrete components such as operational amplifiers, resistors, and Zener diodes to build current sensors. This solution uses the zero-drift amplifier AD8628 as the core device.
- Use high-voltage bidirectional shunt monitors such as AD8210 to improve integration, and use other external devices to expand dynamic range and accuracy.
- Use devices optimized for applications, such as the latest AD8217. The AD8217 is an easy-to-use and highly integrated zero-drift current sensor with an input common-mode voltage range of 4.5V to 80V.
Configure a standard operational amplifier for high-end current detection
Figure 1 shows a discrete solution based on an operational amplifier using AD8628. The same setting is valid when other operational amplifiers are used, but they must have the following characteristics as much as possible: low input offset voltage, low offset voltage drift, low input bias current, and rail-to-rail input and output swing capability. Other recommended amplifiers include AD8538, AD8571, and AD8551.
Figure 1. Discrete high-current detection solution using operational amplifiers
This circuit monitors the high-side currentIThe amplifier is turned on and biased by a Zener diode, which is rated at 5.1V in this example.The use of diodes ensures that the amplifier can work safely at a high common-mode level, and its power supply voltage is stabilized within the allowable power supply limit, while the MOSFET converts its output into current, which is then used by the resistorRL. Converted to a ground-referenced voltage. In this way, the output voltage can be fed to converters, analog processors and other ground-referenced devices (such as operational amplifiers or comparators) for further signal conditioning.
In this configuration,RGVoltage on andRSHUNTThe voltages are equal because the feedback through the MOSFET will keep the two high-impedance inputs of the op amp at the same voltage.go throughRGThe current flows through the FET andRLproduceVOUTPUT.The current flowing through the shunt resistorI,andVOUTPUTThe relationship of can be expressed by Formula 1:
RSHUNTchoose:RSHUNTThe maximum value of is determined by the allowable power dissipation at the maximum current, and the minimum value is determined by the input range and error budget of the operational amplifier. In general, in order to monitor currents above 10A,RSHUNTThe value of is between 1mΩ and 10mΩ.If a single resistor cannot meet the power consumption requirements, or is too large for the PCB, thenRSHUNTIt may have to be composed of multiple resistors in parallel.
RGchoose:RGUsed to convert a current proportional to the high-side current to the low-side.RGThe maximum value of is determined by the drain-source leakage current of the P-channel MOSFET.Assuming that the common P-channel enhancement type vertical DMOS transistor BSS84 is used, then under various conditionsIDSSThe maximum value is shown in Table 1.
Table 1. Drain-source leakage current
|VGS =0V;VDS = –40 V; TJ °C||–100 nA|
|VGS =0V;VDS=–50V;TJ °C||–10 µA|
|VGS = 0 V; VDS = –50 V; TJ = 125°C||–60 µA|
Take LDMOS drain current monitoring as an example, the common mode voltage is 28V,IDSSIt is 100nA.pass throughRLThe mirror image of the minimum current should be at leastIDSS20 times.therefore
RGThe minimum value of is determined by the allowable mirror current power consumption at the maximum load current:
RBIASchoose:pass throughRBIASThe current is shunted to generate the quiescent current of the operational amplifier and the basically constant Zener diode voltageVZ, (It determines the power supply voltage of the operational amplifier).When the amplifier currentISUPPLY, Is actually 0 and,VINWhen it is the maximum value, make sure that the current flowing through the Zener diode does not exceed its maximum regulation currentIZ_MAX:
whenISUPPLY, Is the maximum value andVINWhen it is the minimum value, in order to ensure that the diode voltage is stable, the current flowing through it should be greater than its maximum operating currentIZ_MIN:
Zener diode andRBIASThey are the key components of this solution because they eliminate the high common-mode voltage of subsequent circuits and support the use of low-voltage precision operational amplifiers. In order to maintain the highest voltage stability, the Zener diode should have low dynamic resistance and low temperature drift characteristics.
R1choose:R1Used to limit the amplifier input current when the input transient exceeds the power supply voltage of the operational amplifier. It is recommended to use a 10kΩ resistor.
The offset voltage of the selected op ampVOSAnd offset currentIOSIt is a very important indicator, especially when the shunt resistance and load current are very low.VOS+IOSXR1Must be less thanIMINXRSHUNT, Otherwise the amplifier may be saturated. Therefore, for best performance, it is best to use a rail-to-rail input amplifier with zero crossover distortion.
For this discrete solution, another issue that needs to be considered is temperature drift. Even with a zero-drift amplifier, it is very difficult to optimize or costly to optimize the drift caused by the following discrete components: Zener diodes, MOSFETs, and resistors.It can be seen from Table 1 that whenVGS=0V andVDS=-50V, as the operating temperature changes from 25°C to 125°C, the MOSFET’sIDSSThe maximum value changed from -10μA to -60μA. This drift will reduce the accuracy of the system over the entire temperature range, especially when the monitored current is very low. The drift characteristics of the Zener diode will affect the stability of the amplifier’s power supply, so the amplifier used should have high power supply rejection (PSR) performance.
In addition, the designer must be aware of the low efficiency of this solution becauseRBIASA lot of power is consumed.For example, if the bus common mode voltage is 28V, the Zener diode output voltage is 5.1V andRBIASFor a resistance of 1000Ω, the useless power consumption of the circuit will exceed 0.52W. This will increase the power budget, which must be considered when designing.
Using AD8210 and external devices for high-side current detection
Figure 2a shows a simplified block diagram of the AD8210 integrated high-voltage bidirectional shunt monitor; Figure 2b shows a unidirectional application using an external reference voltage source.
Figure 2. (a) High-voltage bidirectional shunt monitor AD8210
(b) Wide-range unidirectional applications using external reference voltage sources
AD8210 can amplify the small differential input voltage generated when positive or negative current flows through the shunt resistor, while suppressing high common-mode voltage (up to 65V), and provides a ground-referenced buffered output.
As shown in Figure 2a, it mainly includes two modules: a differential amplifier and an instrumentation amplifier.Input throughR1withR2Connect to differential amplifier A1. A1 uses Q1 and Q2 to adjust the flow throughR1withR2The small current makes the voltage on its own input terminal zero. When the input signal of AD8210 is 0V,R1withR2The currents in are equal. When the differential signal is non-zero, the current in one resistor increases, and the current in the other resistor decreases. The current difference is proportional to the size and polarity of the input signal.
R3withR4Convert the differential current flowing through Q1 and Q2 into a differential voltage. A2 is configured as an instrumentation amplifier to convert this differential voltage to a single-ended output voltage. The gain is set to 20V/V internally through finely adjusted thin film resistors.
useVREF1withVREF2The pin can easily adjust the output reference voltage. In a typical configuration that handles bidirectional current,VREF1Connected toVCC,andVREF2Connect to GND.In this case, when the input signal is 0V, the output isVCC/2 is the center voltage. Therefore, for a 5V power supply, the output is centered at 2.5V. Depending on the direction of the current on the shunt resistor, the output will be greater or less than 2.5V.
This configuration is very suitable for charging/discharging applications, but if the user needs to use the entire output range to measure a unidirectional current, then a typical method is to use an external source to set the range, as shown in Figure 2b.At this time, a resistor divider is buffered by an operational amplifier to drive the connectedVREF1withVREF2Pin to offset the output.
When the load current is close to zero, it is difficult to monitor the load current by the amplifier alone. When using a 5V power supply, the linear output range of the AD8210 is 50mV to 4.9V. Assuming that the shunt resistance in the application is 10mΩ, the minimum current flowing through it must be greater than 250mA to ensure that the output of the AD8210 is higher than its lowest point of 50mV.
The configuration shown in Figure 2b introduces an offset in order to measure smaller currents. When the amplifier gain is 20V/V, the relationship between the output voltage and the monitored current can be expressed by the formula 2 table:
For example, when the resistanceR1withR2When they are 9800Ω and 200Ω respectively, the offset voltage is 100mV. When the differential input is 0V, the output of AD8210 is 100mV, which still falls within the linear range.If the shunt current range is 50mA to 20A, whenRSUPPLY=10mΩ, the input range will be 0.5mV to 200mV, the output range of AD8210 is 10mV to 4V plus the offset voltage, that is, 0.11V to 4.1V, which is completely within its rated linear range.
In fact, with this configuration, designers can offset the output of the AD8210 to any point within the power supply range to handle any current range with any asymmetry. Since the precision-adjusted resistor is internally connected to the reference input, an operational amplifier is needed to buffer the voltage divider. For best results, these inputs should be driven with low impedance. Precision low-cost operational amplifiers that can be used to buffer external reference voltage sources include AD8541, AD8601, AD8603, AD8605, AD8613, AD8691, and AD8655.
Compared with discrete solutions, this integrated solution requires the shunt monitor to have a high common-mode voltage range. When the output voltage range cannot meet the current detection range requirements, it also requires an output offset. But it can handle bidirectional current monitoring and avoid the temperature drift and power consumption problems mentioned above. The guaranteed maximum offset drift and gain drift of the AD8210 are 8μV/°C and 20ppm/°C, respectively. If AD8603 is used as a buffer, the offset contributed by it is only 1µV/°C, which is negligible compared to the already low offset voltage drift of AD8210.Voltage dividerR1withR2The power consumption is:
Calculating with the parameters shown in Figure 2b, its power consumption is only 1.2mW.
Using the zero-drift AD8217 for high-side current monitoring
ADI recently introduced a high-voltage current sensor AD8217, which has zero drift and 500kHz bandwidth, specifically used to enhance the resolution and accuracy of wide temperature, input common mode and differential voltage range. Figure 3a shows a simplified block diagram of the device; Figure 3b shows a typical application.
Figure 3. (a) High-resolution, zero-drift shunt monitor AD8217
(b) Use AD8217 for high-end current detection
In order to measure the very small current flowing through a small shunt resistor, the AD8217 provides a minimum output range of 20mV (over the entire temperature range), which is better than the 50mV range of the AD8210. Therefore, if the minimum load current monitored on the shunt resistor produces a minimum output of 20mV in the current sensor (equivalent to a minimum input of 1mV), the user can choose to configure the AD8217 as shown in Figure 3b. The relationship between the output voltage of the AD8217 and the input current can be expressed by formula 3:
AD8217 has a built-in low dropout regulator (LDO), which can provide a constant voltage power supply for the amplifier. The LDO can withstand high common-mode voltages from 4.5V to 80V, and its function is basically similar to the Zener diode in Figure 1.
The factory setting gain of AD8217 is 20V/V, and the maximum gain error in the whole temperature range is ±0.35%. The initial offset rating over the entire temperature range is ±300µV, and the temperature drift is very small, only ±100nV/°C. These features can improve any error budget. The buffered output voltage can be directly interfaced with any typical analog-to-digital converter. When the input differential voltage is at least 1mV, the AD8217 can provide the correct output voltage regardless of whether there is a common-mode voltage. When using a 10mΩ shunt resistor as in the previous example, the minimum current can be as low as 100mA.
The single-chip solution avoids the temperature drift and power consumption issues of discrete solutions.
The following section will give the test results obtained by comparing these three different methods. During the test, the input current flowing through the shunt resistor is adjusted by changing the input voltage and load resistance. In the data shown, an initial calibration has been performed to eliminate the initial gain and offset errors associated with all devices on the board.
Figure 4 is measured using the circuit shown in Figure 1RLThe output voltage on theRSUPPLYThe linear relationship graph between the low-end value of the input current.RSUPPLY10mΩ;RGIs 13Ω;RBIASIs 100Ω;R110kΩ; load resistance is 200Ω;RLIt is 200Ω; Zener diode output is 5.1V; operational amplifier is AD8628; MOSFET is BSS84. The maximum relative error is 0.69%, and the average error after calibration is 0.21%.
Figure 4. Low current test results obtained with AD8628 in Figure 1
Figure 5 shows the output voltage and flow of AD8210 measured by the circuit shown in Figure 2b.RSUPPLYThe linear relationship graph between the low-end value of the input current.RSUPPLY10mΩ;R120kΩ;R2It is 0.5kΩ; the load resistance is 200Ω; the external reference voltage buffer is AD8603. The maximum relative error is 0.03%, and the average error after calibration is 0.01%.
Figure 5. Low current test results obtained with AD8210 in Figure 2b
Figure 6 shows the output voltage and flow of AD8217 measured by the circuit shown in Figure 3b.RSUPPLYThe linear relationship graph between the low-end value of the input current.RSUPPLYIt is 10mΩ, and the load resistance is 50Ω. The maximum relative error is 0.088%, and the average error after linear correction is 0.025%.
Figure 6. Low current test results obtained by using AD8217 in Figure 3b
Note that testing must focus on the low end of the range, rather than covering the entire range of 50mA to 20A. The reason is that the linearity change is mainly in the low output voltage (low unipolar current) part of the range.
In addition, temperature experiments were performed on each solution at –40°C, +25°C, and +85°C. Table 2 shows the maximum relative error and average error when calibrating data at –40°C and +85°C using the correction coefficient at +25°C.
Table 2. Maximum error and average error when using the same correction factor at different temperatures
|–40°C||Maximum error (%)||11.982||2.117||0.271|
|+25°C||Maximum error (%)||1.806||0.075||0.103|
|+85°C||Maximum error (%)||6.632||3.800||0.918|
If a temperature sensor is available in the system, different correction coefficients can be used to calibrate the data at different temperatures, but this will result in an increase in the number of devices and an increase in manufacturing costs. Table 3 shows the maximum relative error and average error when using different correction coefficients at –40°C, +25°C, and +85°C.
Table 3. Maximum error and average error when using different correction coefficients at different temperatures
|–40°C||Maximum error (%)||1.981||0.022||0.114|
|+25°C||Maximum error (%)||1.806||0.075||0.103|
|+85°C||Maximum error (%)||1.844||0.038||0.075|
Temperature experiments show that devices using auto-zero technology can provide high-precision performance in a wide temperature range, especially the AD8217.
Figure 7. Temperature experiment of AD8628 discrete solution
Figure 8. Temperature experiment of AD8210 integrated solution
Figure 9. Temperature experiment of AD8217 single-chip solution
The test results show that all three solutions can be used for high-end current detection with a wide dynamic range: the output of all three solutions is linear, and the solution using AD8217 has the best error performance and does not need to be independent power supply. In addition, the offset drift characteristic of ±100-nV/°C makes it very suitable for use in the temperature range of –40°C to +125°C, and can provide the highest accuracy performance in the temperature range. As far as system design is concerned, single-chip solutions can save PCB area, simplify PCB layout, reduce system costs, and improve reliability. These features are particularly suitable for unidirectional current sensing applications where the load current range is wide and the dynamic range is critical.
According to the test results, it can be known that the AD8217 is the most suitable one of the three solutions for unidirectional high-end current detection and monitoring applications with a wide dynamic range. We also noticed that the working range of the AD8210 solution can be as low as 0V input, which may be beneficial for detecting short-to-ground conditions. It should also be noted that the AD8210 can monitor bipolar currents in a single chip, for example in charge/discharge applications.
In actual system design that requires the best system performance, calibration and temperature detection are recommended.
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